Controller, motor control system having the controller, and electric power steering system having the motor control system

ABSTRACT

A technique for reducing an operation noise even when a high-pass filter is used for controlling a motor is provided. A controller is used in a motor control system for driving a motor by using a drive circuit and an inverter. The controller includes a current control block. The controller performs feedforward control by using a current value, compensates for a term of a self-inductance of the motor included in a current control block by an inverse model, compensates for a phase characteristic of a transfer function of the inverse model by an advance component, and corrects a gain characteristic of the transfer function of the inverse model by a function of a physical quantity obtained based on an angular velocity of the motor, thereby compensating for a phase delay and a gain reduction of a torque output generated by the self-inductance.

CROSS-REFERENCE TO RELATED APPLICATIONS

This is the U.S. national stage of application No. PCT/JP2018/043211,filed on Nov. 22, 2018, and the present invention claims priority under35 U.S.C. § 119 to Japanese Application No. 2017-234682 filed on Dec. 6,2017.

FIELD OF THE INVENTION

The present disclosure relates to a controller, a motor control systemhaving the controller, and an electric power steering system having themotor control system.

BACKGROUND

Japanese Unexamined Patent Application Publication No. 2017-034760discloses a technique for controlling by using an inverse model of afeedforward technique has been known.

In motor current control of an electric power steering, feedback controlis commonly used. However, when the feedback control is performed, thereis a problem that an unpleasant motor operation noise is generated inresponse to a noise included in a current detector. In order to suppressthe generation of the unpleasant motor operation noise, a method whichdoes not directly use a current detection value, is conceivable.However, when the current detection value is not directly used, there isa problem that motor output torque varies due to parameter variations ofthe motor and motor drive circuit characteristics in the existingfeedforward control technique.

In the existing technique described above, a d-axis actual current, aq-axis actual current, and a zero-phase actual current are passedthrough a high-pass filter. However, when the high-pass filter is used,sensitivity of a torque command value noise is increased, so that themotor operation noise is increased.

SUMMARY

An exemplary controller of the present disclosure is a controller usedin a motor control system for driving a motor by using a drive circuitand an inverter, and includes a current control block, in whichfeedforward control based on a current value is performed, a term of aself-inductance of the motor included in the current control block iscompensated by an inverse model, further a phase characteristic of atransfer function of the inverse model is compensated by an advancecomponent, and a phase delay and a gain reduction of a torque outputgenerated by the self-inductance is compensated by correcting a gaincharacteristic of the transfer function of the inverse model by afunction of a physical quantity obtained based on an angular velocity ofthe motor.

Advantageous Effects of Invention

According to an exemplary embodiment of the present disclosure,feedforward control based on a current value is performed, and a term ofa self-inductance of a motor included in a current control block iscompensated by an inverse model. In addition, a phase characteristic ofa transfer function of the inverse model is compensated by an advancecomponent, and a gain characteristic is corrected by a function of aphysical quantity obtained based on an angular velocity of the motor.Since a phase delay and a gain reduction of a torque output caused bythe self-inductance can be compensated, even when a high-pass filter isused for controlling the motor, an operation noise can be reduced.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a schematic diagram of a hardware block of a motor controlsystem 1000 according to an exemplary embodiment of the presentdisclosure;

FIG. 2 schematically illustrates a hardware configuration of an inverter300 in the motor control system 1000 according to the presentembodiment;

FIG. 3 is a block diagram illustrating an internal configuration of acontroller 100;

FIG. 4 is a control block diagram illustrating details of an exemplaryU-phase processing circuit 104 a of the present disclosure;

FIG. 5 is a diagram illustrating a configuration of a U-phase processingcircuit 104 a according to a modification; and

FIG. 6 schematically illustrates a typical configuration of an EPSsystem 2000 according to an exemplary embodiment.

DETAILED DESCRIPTION

Hereinafter, embodiments of a controller, a motor control system havingthe controller, and an electric power steering system having the motorcontrol system of the present disclosure will be described in detailwith reference to the accompanying drawings. However, in order to avoidthe following description from being unnecessarily redundant and tofacilitate understanding of those skilled in the art, an unnecessarydetailed description may be omitted. For example, a detailed descriptionof the matters already well known and a duplicate description ofsubstantially the same configuration may be omitted.

In the embodiments and modifications of the present disclosure, aconfiguration for achieving the purpose described in each of thefollowing items will be described. An outline of the configuration willalso be described.

(1) Reduction of Motor Current Detection Noise Sensitivity

A feedforward (FF) type is applied as basic control. In the embodiment,feedback (FB) control using a current detection value is not performed.Parameter variations which cause a problem with the FF type control iscorrected by an observer using a current value. Combining withthree-phase independent control produces the effect described above.

(2) Reduction of Torque Command Value Noise Sensitivity

In performing the FF type control, a motor self-inductance iscompensated by an inverse model. In this case, a high-pass filterincreases the noise sensitivity. In fact, there may be cases where amotor operation noise increases due to a noise sensitivity problem. Thepresent inventor has created a method for reducing such noisesensitivity.

(3) Compensation of a Drive System Non-Linear Element

The non-linear element having the largest influence on the operationnoise is a dead band of a drive circuit. The dead band occurs when thecurrent crosses zero. The present inventor estimates a timing at whichthe current crosses zero, and performs dead band compensation using theestimation result.

Hereinafter, the embodiments of the present disclosure will bedescribed.

A current controller capable of reducing the motor current detectionnoise sensitivity will be described below.

FIG. 1 schematically illustrates a hardware block of a motor controlsystem 1000 according to an exemplary embodiment of the presentdisclosure.

The motor control system 1000 typically includes a motor M, a controller(control circuit) 100, a drive circuit 200, an inverter (also referredto as an “inverter circuit”) 300, a plurality of current sensors 400, ananalog/digital conversion circuit (hereinafter referred to as an “ADconverter”) 500, a Read Only Memory (ROM) 600, and a position sensor700. The motor control system 1000 is modularized and manufactured, forexample, as a motor module having a motor, a sensor, a driver, and acontroller. In the description, the motor control system 1000 will bedescribed by taking an example of a system having the motor M as aconstituent element. However, the motor control system 1000 may be asystem for driving the motor M, which does not include the motor M as aconstituent element.

The motor M is a surface permanent magnet type (SPM) motor, and is, forexample, a surface permanent magnet type synchronous motor (SPMSM). Themotor M includes, for example, three-phase (U-phase, V-phase, andW-phase) windings (not illustrated). The three-phase windings areelectrically connected to the inverter 300. The present disclosure isnot limited to the three-phase motors, but also includes multi-phasemotors such as five-phase, seven-phase motors and the like. In thedescription, the embodiment of the present disclosure will be describedusing a motor control system that controls the three-phase motor as anexample. As the motor M, a motor having a relatively small mutualinductance between phases, for example, a motor having 10 poles and 12slots, and a motor having 14 poles and 12 slots may be used.

The controller 100 is, for example, a micro control unit (MCU). Thecontroller 100 controls the entire motor control system 1000 andcontrols torque and a rotational speed of the motor M by, for example,vector control. The motor M can be controlled not only by the vectorcontrol but also by other closed loop control. The rotational speed isrepresented by a rotational speed (rpm) at which a rotor rotates perunit time (for example, per one minute) or a rotational speed (rps) atwhich the rotor rotates per unit time (for example, per one second). Thevector control is a method of decomposing a current flowing through amotor into a current component contributing to generation of torque anda current component contributing to generation of magnetic flux, andindependently controlling each current component orthogonal to eachother. The controller 100 sets target current values according to, forexample, actual current values measured by the plurality of currentsensors 400, a rotor angle estimated based on the actual current values,and the like. The controller 100 generates Pulse Width Modulation (PWM)signals based on the target current values, and outputs the PWM signalsto the drive circuit 200.

The drive circuit 200 is, for example, a gate driver. The drive circuit200 generates control signals for controlling switching operation ofswitching elements in the inverter 300 according to the PWM signalsoutput from the controller 100. The drive circuit 200 may be mounted onthe controller 100.

The inverter 300 converts DC power supplied from, for example, a DCpower supply (not illustrated) into AC power, and drives the motor Mwith the converted AC power. For example, the inverter 300 converts theDC power into three-phase AC power, which is a pseudo sine wave of theU-phase, the V-phase, and the W-phase, based on the control signalsoutput from the drive circuit 200. The motor M is driven by theconverted three-phase AC power.

The plurality of current sensors 400 includes at least two currentsensors that detect at least two currents among the currents flowingthrough the U-phase, V-phase, and W-phase windings of the motor M. Inthe present embodiment, the plurality of current sensors 400 includestwo current sensors 400A and 400B (see FIG. 2) for detecting currentsflowing through the U-phase and the V-phase. Naturally, the plurality ofcurrent sensors 400 may have three current sensors that detect threecurrents flowing through the U-phase, V-phase, and W-phase windings, or,for example, may have two current sensors that detect currents flowingthrough the V-phase and W-phase, or currents flowing through the W-phaseand U-phase. The current sensor includes, for example, a shunt resistorand a current detection circuit (not illustrated) for detecting acurrent flowing through the shunt resistor. The resistance value of theshunt resistor is, for example, about 0.1Ω.

The AD converter 500 samples analog signals output from the plurality ofcurrent sensors 400, converts the analog signals into digital signals,and outputs the converted digital signals to the controller 100. Thecontroller 100 may perform AD conversion. In this case, the controller100 receives the detected current signals (analog signals) directly fromthe plurality of current sensors 400.

The ROM 600 is, for example, a writable memory (for example, PROM), arewritable memory (for example, flash memory), or a read-only memory.The ROM 600 stores a control program having a command group for causingthe controller 100 to control the motor M. For example, the controlprogram is temporarily loaded on a RAM (not illustrated) at the time ofbooting. The ROM 600 does not need to be externally attached to thecontroller 100, and may be mounted on the controller 100. The controller100 on which the ROM 600 is mounted may be, for example, the MCUdescribed above.

The position sensor 700 is, the position sensor 700 is disposed on themotor M, detects the rotor angle P, and outputs the detected rotor angleP to the controller 100. The position sensor 700 may be implemented by,for example, a combination of a magnetoresistive (MR) sensor having anMR element and a sensor magnet. The position sensor 700 may be alsoimplemented by using, for example, a Hall IC including a Hall element ora resolver.

In the exemplary embodiment, the controller 100 is implemented by afield programmable gate array (FPGA) with an incorporated CPU core. Inthe FPGA hardware logic circuit, an observer block, a current controlblock, and a vector control operation block, which will be describedlater, are constructed. For example, the CPU core calculates a torquecommand value by software processing. Each block in the FPGA generates aPWM signal using the torque command value (T_(ref)) received from theCPU core and a rotor rotational position of the motor M measured by theposition sensor 700, that is, the rotor angle (P), a current measurementvalue (I_(a), I_(b), I_(c)) received from the AD converter 500, and thelike.

In FIG. 1, the inverter 300 is configured by one system, but may beconfigured by a plurality of systems, for example, two systems. Evenwith the plurality of systems, a controller having the same orequivalent function and configuration as the controller 100 may beemployed for each of the plurality of systems, or a different controllermay be employed.

Each constituent element configuring the motor control system 1000illustrated in FIG. 1, for example, the motor M, the controller 100, thedrive circuit 200, the inverter 300, and the like may be integrallyhoused in a housing (not illustrated). Such a configuration ismanufactured and sold as a so-called “mechanically/electricallyintegrated motor”. In the mechanically/electrically integrated motor,since the various constituent elements are housed in the housing, thereis no need to design an arrangement of each constituent element, aninstallation space, and a wiring layout. As a result, space saving ofthe motor and its peripheral circuit, and simplification of the designcan be realized. The controller 100 according to the present embodimentcan suppress the operation noise generated by the rotation of the motorM by using a feedforward control technique described later. Byintegrating the controller 100 and the motor M, it is possible toprovide a space-saving and low-noise “mechanically/electricallyintegrated motor”. The “mechanically/electrically integrated motor” mayfurther include the current sensor 400, the converter 500, and the ROM600.

With reference to FIG. 2, a hardware configuration of the inverter 300will be described in detail.

FIG. 2 schematically illustrates the hardware configuration of theinverter 300 in the motor control system 1000 according to the presentembodiment.

The inverter 300 includes three low-side switching elements and threehigh-side switching elements. Illustrated switching elements SW_L1,SW_L2, and SW_L3 are low-side switching elements, and switching elementsSW_H1, SW_H2, and SW_H3 are high-side switching elements. As theswitching element, for example, a semiconductor switching element suchas a field effect transistor (FET, typically MOSFET), an insulated gatebipolar transistor (IGBT), or the like can be used. The switchingelement includes a reflux diode for flowing a regenerative currentflowing toward the motor M.

FIG. 2 illustrates shunt resistors Rs of three current sensors 400A,400B, and 400C for detecting currents flowing through the U-phase,V-phase, and W-phase. As illustrated, for example, the shunt resistor Rsmay be electrically connected between the low-side switching element andthe ground. Alternatively, for example, the shunt resistor Rs may beelectrically connected between the high-side switching element and apower supply.

The controller 100 can drive the motor M by performing, for example,control by three-phase energization (hereinafter, referred to as“three-phase energization control”) based on the vector control. Forexample, the controller 100 generates PWM signals for performing thethree-phase energization control, and outputs the PWM signals to thedrive circuit 200. The drive circuit 200 generates a gate control signalfor controlling the switching operation of each FET in the inverter 300based on the PWM signal, and supplies the gate control signal to thegate of each FET.

Although three current sensors 400A, 400B, and 400C are provided in FIG.2, the number of current sensors may be two. For example, the currentsensor 400C for detecting the current flowing through the W-phase can beomitted. In this case, the current flowing through the W-phase may bedetected by calculation instead of measurement. In the three-phaseenergization control, the sum of the currents flowing through respectivephases becomes ideally zero. When the currents flowing through theU-phase and V-phase are detected by the current sensors 400A and 400B,respectively, a value obtained by inverting a sign of the sum of theU-phase current and the V-phase current can be calculated as the currentvalue flowing through the W-phase.

According to the present disclosure, three current sensors may beprovided to detect the current flowing through each of the three phases,or two current sensors may be provided to detect the currents in the twophases and the current flowing through the remaining one phase may becalculated by performing the above-described calculation.

FIG. 3 is a block diagram illustrating an internal configuration of thecontroller 100. The controller 100 includes current controllers 102 a,102 b, and 102 c, and a voltage-duty converter 180. The currentcontroller 102 a receives a torque command value T_(refa) and a U-phasecurrent value I_(a), and outputs a command voltage V_(refa). The currentcontroller 102 b receives a torque command value T_(refb) and a U-phasecurrent value I_(b), and outputs a command voltage V_(refb). The currentcontroller 102 c receives a torque command value T_(refc) and a U-phasecurrent value I_(c), and outputs a command voltage V_(refc).

In the description, a description will be given assuming that the threecomponents T_(refa), T_(refb), and T_(refc) of the torque command valueT_(ref) are given values. Each of these values is generated by, forexample, a CPU core (not illustrated) of the controller 100. Sinceprocessing for generating the torque command value is well known, adescription thereof will be omitted.

The voltage-duty converter 180 performs a voltage-duty conversion. Thevoltage-duty conversion is processing for generating a PWM signal from acommand voltage. The PWM signal represents a voltage command value.Specifically, the voltage-duty converter 180 generates a PWM signalV_(dutya) from the command voltage V_(refa). Similarly, the voltage-dutyconverter 180 generates PWM signals V_(dutyb) and V_(dutyc) from thecommand voltages V_(refb) and V_(refc), respectively. Since thevoltage-duty conversion is well known, a detailed description thereofwill be omitted in the description.

Next, details of the current controllers 102 a to 102 c will bedescribed. Hereinafter, a U-phase processing circuit 104 a including thecurrent controller 102 a and the voltage-duty converter 180 will bedescribed as an example. Since both the current controller 102 b and thecurrent controller 102 c are the same, illustration and a descriptionthereof are omitted.

FIG. 4 is a control block diagram illustrating details of the U-phaseprocessing circuit 104 a. A portion excluding the voltage-duty converter180 in the U-phase processing circuit 104 a corresponds to the currentcontroller 102 a (FIG. 3).

The U-phase processing circuit 104 includes a torque/current conversionblock 110 a, a current control block 120 a, an adaptive control block130 a, and an adder 140 a. The respective blocks and the adder representarithmetic operations. Therefore, a “block” can also be read as an“operation”. All operations may be implemented by the hardware logic ofthe FPGA, or one or more operations may be implemented by one or morearithmetic circuits.

The torque/current conversion block 110 a converts the torque commandvalue T_(refa) into a current command value I_(refa).

The current control block 120 a and the adder 140 a are operation blockscorresponding to an operation of a voltage equation to be describedlater. The current control block 120 a functions as a high-pass filter.The current control block 120 a sequentially corrects a resistance valueR_(tha) by a modeling error ΔR_(tha) calculated by the adaptive controlblock 130 a. That is, a voltage value is obtained by using the previousresistance value R_(tha)+ΔR_(tha) as a new R_(tha).

The adaptive control block 130 a outputs the modeling error ΔR_(tha) byusing the current value I_(a) flowing through the U-phase. The adaptivecontrol block 130 a has a first operation block 132 a that performs thesame operation as the current control block 120 a, and a secondoperation block 134 a. Of these, the latter second operation block 134 afunctions as an “observer”. Hereinafter, the second operation block 134a will be described as an “observer block 134 a”. As is apparent fromthe description of the observer block 134 a in FIG. 4, the observer is aprimary low-pass filter with a time constant T₁.

Note that the first operation block 132 a includes a differential symbol“d/dt”, and is represented in a time domain, while the observer block134 a is represented in an s domain using a variable “s”. The reason forusing the variable s is to clarify that the observer is the primarylow-pass filter with the time constant T₁. It should be noted that it isfor convenience of understanding.

A signal input to the adaptive control block 130 a (signal to befiltered) is not a white noise but a colored noise. In the presentembodiment, the adaptive control block 130 a does not perform afiltering operation using the least squares method.

Hereinafter, before describing a meaning of each block illustrated inFIG. 4, how the operation corresponding to each block is derived will bedescribed.

When the motor is rotating at an angular velocity ω, power injected intoa coil is E·I, and power generated by the coil is T·ω. Where E isvoltage, I is current, and T is torque.

According to the energy conservation law, a following equation (1)holds.EI=Tω  (1)

A following equation (2) is obtained by modifying the equation (1).T−EI/ω  (2)

The present inventor has considered to avoid using the current I becauseit contains much noise. When the current I is obtained from a voltageequation (equation (3)) and substituted into the equation (1), anequation (4) is obtained.I=f(V)  (3)T=(E/ω)f(V)  (4)

Here, the voltage equation is shown in a following equation (5).

$\begin{matrix}{\begin{bmatrix}v_{a} \\v_{b} \\v_{c}\end{bmatrix} = {{R_{th}\begin{bmatrix}i_{a} \\i_{b} \\i_{c}\end{bmatrix}} + {\begin{bmatrix}L_{p\; h} & M & M \\M & L_{p\; h} & M \\M & M & L_{p\; h}\end{bmatrix}\frac{d}{dt}( \begin{bmatrix}i_{a} \\i_{b} \\i_{c}\end{bmatrix} )} + \begin{bmatrix}e_{a} \\e_{b} \\e_{c}\end{bmatrix}}} & (5)\end{matrix}$

When the vector on the left side is represented by V, the vector commonto the first and second terms on the right side is represented by I, thematrix of the second term on the right side is represented by L, and thevector of the third term on the right side is represented by E, afollowing equation (6) is obtained and an equation (7) is obtained byfurther being modified.

$\begin{matrix}{V = {{R_{th}I} + {L\;\frac{d}{dt}I} + E}} & (6) \\{I = {( {R_{th} + {L\;\frac{d}{dt}}} )^{- 1}( {V - E} )}} & (7)\end{matrix}$

Here, the inductance is generalized and represented as shown in anequation (8).

$\begin{matrix}{\mspace{20mu}{{\lbrack L\rbrack = {l_{s} + \begin{bmatrix}L_{u} & M_{uv} & M_{uw} \\M_{vu} & L_{v} & M_{vw} \\M_{wu} & M_{wv} & L_{w}\end{bmatrix}}}\mspace{20mu}{L_{u} = {\sum\limits_{k = 0}^{\infty}{L_{k}\;\cos\; 2k\;\theta}}}\mspace{20mu}{L_{v} = {\sum\limits_{k = 0}^{\infty}{L_{k}\cos\; 2k\;( {\theta - \frac{2\pi}{3}} )}}}\mspace{20mu}{L_{w} = {\sum\limits_{k = 0}^{\infty}{L_{k}\cos\; 2k\;( {\theta - \frac{4\pi}{3}} )}}}\mspace{20mu}{M_{uv} = {M_{vu} = {\sum\limits_{k = 0}^{\infty}{M_{k}\cos\; 2{k( {\theta - \frac{4}{3}} )}}}}}\mspace{20mu}{M_{vw} = {M_{uv} = {\sum\limits_{k = 0}^{\infty}{L_{k}\cos\; 2k\;(\theta)}}}}\mspace{20mu}{M_{wu} = {M_{uw} = {\sum\limits_{k = 0}^{\infty}{L_{k}\cos\; 2{k( {\theta - \frac{2\pi}{3}} )}}}}}}} & (8)\end{matrix}$

Components of the inductance up to a sixth order are exemplified in afollowing equation (9). As described later, in the present embodiment,feedforward (FF) type control (FF control) is performed. When performingthe FF control, the higher-order components of the inductance disappear.

$\begin{matrix}{\mspace{20mu}{\lbrack L\rbrack = {{l_{s}\begin{bmatrix}0 & 1 & 0 \\0 & 1 & 0 \\0 & 0 & 1\end{bmatrix}} + {L_{g\; 0}\begin{bmatrix}1 & {- \frac{1}{2}} & {- \frac{1}{2}} \\{- \frac{1}{2}} & 1 & {- \frac{1}{2}} \\{- \frac{1}{2}} & {- \frac{1}{2}} & 1\end{bmatrix}}\mspace{20mu} - {L_{g\; 2}\begin{bmatrix}{\cos\; 2\theta} & {\cos\; 2( {\theta - \frac{4\pi}{3}} )} & {\cos\; 2( {\theta - \frac{2\pi}{3}} )} \\{\cos\; 2( {\theta - \frac{4\pi}{3}} )} & {\cos\; 2( {\theta - \frac{2\pi}{3}} )} & {\cos\; 2\theta} \\{\cos\; 2( {\theta - \frac{2\pi}{3}} )} & {\cos\; 2\theta} & {\cos\; 2( {\theta - \frac{4\pi}{3}} )}\end{bmatrix}}\mspace{20mu} - {L_{g\; 4}\begin{bmatrix}{\cos\; 4\theta} & {\cos\; 4( {\theta - \frac{4\pi}{3}} )} & {\cos\; 4( {\theta - \frac{2\pi}{3}} )} \\{\cos\; 4( {\theta - \frac{4\pi}{3}} )} & {\cos\; 4( {\theta - \frac{2\pi}{3}} )} & {\cos\; 4\theta} \\{\cos\; 4( {\theta - \frac{2\pi}{3}} )} & {\cos\; 4\theta} & {\cos\; 4( {\theta - \frac{4\pi}{3}} )}\end{bmatrix}}\mspace{20mu} - {L_{g\; 6}\begin{bmatrix}{\cos\; 6\theta} & {\cos\; 6( {\theta - \frac{4\pi}{3}} )} & {\cos\; 6( {\theta - \frac{2\pi}{3}} )} \\{\cos\; 6( {\theta - \frac{4\pi}{3}} )} & {\cos\; 6( {\theta - \frac{2\pi}{3}} )} & {\cos\; 6\theta} \\{\cos\; 6( {\theta - \frac{2\pi}{3}} )} & {\cos\; 6\theta} & {\cos\; 6( {\theta - \frac{4\pi}{3}} )}\end{bmatrix}}}}} & (9)\end{matrix}$

An equation (10) shows E in the equations (6) and (7), taking up to athird harmonic.

$\begin{matrix}{\mspace{20mu}{E = {\begin{bmatrix}e_{a} \\e_{b} \\e_{c}\end{bmatrix} = {{\omega\;{\psi_{1}\begin{bmatrix}{\cos\;\theta} \\{\cos( {\theta - \frac{2\pi}{3}} )} \\{\cos( {\theta + \frac{2\pi}{3}} )}\end{bmatrix}}} + {\omega\;{\psi_{3}\begin{bmatrix}{\cos\; 3\theta} \\{\cos\; 3\theta} \\{\cos\; 3\theta}\end{bmatrix}}}}}}} & (10)\end{matrix}$

From the above, the torque equation is obtained as an equation (11).

$\begin{matrix}{T = {\frac{E}{\omega}( {R_{th} + {L\;\frac{d}{dt}}} )^{- 1}( {V - E} )}} & (11)\end{matrix}$

Here, since the input to the current controller is T and the output fromthe current controller is V, an equation (12) is obtained by rearrangingthe equation (11).

$\begin{matrix}{V = {{( {R_{th} + {L\;\frac{d}{dt}}} )\frac{\omega}{E}T} + E}} & (12)\end{matrix}$

When performing the feedforward control using the equation (12), thepresent inventor has studied compensating for parameter variations. Instudying the parameters to be compensated, the following assumptions aremade. R_(th): Compensate sequentially. L: Use as a fixed value. Notethat the inductance does not change with temperature.

Since the equation (12) becomes a target interphase voltage when thethree phases are independent of each other, a neutral point voltageV_(N) is obtained and corrected as follows.V _(N)−(V _(a) +V _(b) +V _(c))/3  (13)

The phase voltage V_(an) is obtained by a following equation (14).V _(aN) =V _(a) +V _(N)  (14)

From the above, the torque T, the current I, and the voltage V areobtained by equations (15), (16), and (17), respectively. V_(dutya),V_(dutyb) and V_(dutyc) are obtained by an equation (18).

⁢[ T refa T refb T refc ] = [ sin ⁡ ( θ ) sin ⁡ ( θ + 3 2 ⁢ π ) sin ⁡ ( θ - 32 ⁢ π ) ] · T ref ( 15 ) ⁢ [ I refa I refb I refc ] = [ ω 0 0 0 ω 0 0 0 ] ⁡[ T refa T refb T refc ] ( 16 ) [ V refa V refb V refc ] =   [ R tha + La ⁢ s Ms Ms M ⁢ s R thb + L b ⁢ Ms Ms Ms R thc + L c ⁢ s ] ⁢   [ I refa Irefb I refc ] - [ e a e b e c ] ( 17 ) ⁢ [ V DUTYa V DUTYb V ] = [ V refaV refb V refc ] + [ ( V a + V b + V c ) / 3 ( V a + V b + V c ) / 3 ( Va + V b + V c ) / 3 ] ( 18 )

When performing the feedforward control using the equation (12), thepresent inventor has studied compensating the self-inductance L.Specifically, the present inventor has compensated for theself-inductance L by using an inverse model, and compensated for a phasedelay by an advance component. A calculation of the inverse model isperformed by using an abc axis coordinate system instead of a dq axiscoordinate system.

Here, when the self-inductance L is compensated by the inverse model,the present inventor has found a problem that noise sensitivity isincreased. This is because the processing for compensation becomes ahigh-pass filter, the sensitivity of the torque sensor system to noiseincreases, and as a result, the operation noise deteriorates.

Therefore, the present inventor has performed the feedforward control byusing a value of a current flowing through a motor to perform variouscompensations. Specifically, the term of the self-inductance L of themotor M included in the current control block 120 a is compensated bythe inverse model. Further, a phase characteristic of a transferfunction of the inverse model is compensated by the advance component,and a gain characteristic of the transfer function of the inverse modelis corrected by a function of a physical quantity obtained based on theangular velocity of the motor. As a result, it is possible to compensatefor the phase delay and a gain reduction of a torque output caused bythe self-inductance. When compensating for the phase delay and the gainreduction by using the inverse model, it is not essential to provide adisturbance observer to be described next. The feedforward control ispossible without providing the disturbance observer.

Next, the observer will be described.

The present inventor has considered compensating current controlprocessing with the disturbance observer using the current commandvalue. This is because the parameter variations of the output arecompensated, so that the reduction of the current value noise can berealized.

In the present disclosure, the disturbance observer of an input errormodel is used. When this observer is used, the feedforward model and theobserver model are the same, so that design control is facilitated. Theobserver model is shown in an equation (19).

$\begin{matrix}{V_{ab} = {{( {R_{th} + {\Delta\; R_{th}} + {L\;\frac{d}{dt}}} )I} - E}} & (19)\end{matrix}$

In the equation (19), a modeling error between the real and a plantmodel is represented as ΔR_(th). As a result, a following equation (20)is obtained.

$\begin{matrix}{{\begin{bmatrix}V_{DUTYa} \\V_{DUTYb} \\V_{DUTYc}\end{bmatrix} - \begin{bmatrix}V_{aba} \\V_{abb} \\V_{abc}\end{bmatrix}} = \begin{bmatrix}{\Delta\; R_{tha}l_{a}} \\{\Delta\; R_{thb}l_{b}} \\{\Delta\; R_{thc}l_{c}}\end{bmatrix}} & (20)\end{matrix}$

Accordingly, ΔR_(tha), ΔR_(thb), ΔR_(thc) can be obtained by dividingeach component on the right side of the equation (20) by each componentof the detected current I=(I_(a), I_(h), I_(c)).

In the equation (20), the V_(DUTYa) to V_(DUTYc) on the left side arethe respective voltage command values of the PWM signals for the U, V,and W-phases of the voltage-duty converter 180.

In implementation, the noise sensitivity to ΔR estimation is considered,the ΔR_(th) is determined based on a noise-processed signal, and aninternal model of the feedforward controller is adapted. That is, acommon simple adaptive control system is configured. In this case, sincea control target satisfies a condition to be strictly proper, stabilityof the adaptive control system is guaranteed.

Although the above equation is a representation in the time domain, itcan be converted to a representation in the s domain by performingLaplace transform on both sides. In the representation of the s domain,a differential element is replaced by “s”. As a result, the controlblocks and the coupling relationship between the control blocksillustrated in FIG. 4 is represented by the above equation.

When the current value falls below a constant value, for example, whenthe current value falls within a range of zero±a threshold value, theobserver block 134 a may calculate using the previous compensationvalue. When the current value becomes zero or substantially zero, thevoltage is saturated, so that the observer block 134 a cannot estimatethe disturbance R_(th). Accordingly, the compensation can be performednormally by using the previous compensation value when the value becomesa constant value near zero.

Next, a modification of the exemplary embodiment of the presentdisclosure will be described.

FIG. 5 illustrates a configuration of a U-phase processing circuit 104 aaccording to the modification. A configuration of the U-phase processingcircuit illustrated in FIG. 5 differs from the configuration of theU-phase processing circuit illustrated in FIG. 4 in that a dead bandcompensation block 150 a and an adder 160 a are added. Otherconfigurations and operations are the same. Accordingly, the dead bandcompensation block 150 a and the adder 160 a will be described below.For the description of the other components, the description so far iscited.

A “dead band” as referred to hereinafter means a time period in which acurrent cannot flow even though the current is intended to flow. Thedead band is a concept including a time at which no current flows, thatis, a dead time at which the current value is zero, and a period atwhich the current value is rising or is falling from zero. The latter“period” refers to a time period in which substantially the current isconsidered to be substantially zero. The “dead band” arises from arelationship between a non-linear element of a drive system andelectromagnetic compatibility (EMC). The EMC is ability of an apparatusor system to provide no electromagnetic interference wave to interferewith the operation of device or the like to anything, and to withstandinterference from an electromagnetic environment so as to satisfactorilyfunction. The non-linear element of the drive system in the presentembodiment indicates the dead band set for preventing an arm shortcircuit.

Here, a case where the motor M is driven by an electric power steeringsystem will be considered as an example. When a torque ripple isgenerated in the motor M, a driver feels sound or vibration. Forexample, assuming that an output of the motor M is 80 Nm, a person feelssound or vibration unless the torque ripple is set to be less than 0.2Nm. Such a quantization noise becomes a significant problem inapplications requiring accuracy, such as an electric power steering.Therefore, in the electric power steering system, it is required toappropriately compensate for a response of the non-linear elements ofthe drive system, such as the observer block 134 a, the drive circuit200, and the like, so as to reduce the vibration and the operation noiseas much as possible. The present inventor has studied to performcompensation of the non-linear elements of the drive system inconsideration of the dead band.

In the present modification, the dead band compensation block 150 acalculates the compensation value of the non-linear element of the drivesystem based on the dead band compensation value. The dead band of themotor drive circuit is generated at the zero crossing point of thecurrent. The dead band compensation block 150 a outputs a duty valuecorresponding to the dead band at the timing at which the motor currentcrosses zero. The “duty value corresponding to the dead band” may befixed, or may be varied under a predetermined condition.

The adder 160 a adds the duty value at the timing at which the motorcurrent crosses zero, and the duty value corresponding to the dead band.Accordingly, it is possible to control with fewer parameters withrespect to the feedforward control while realizing a low operationnoise.

A timing at which the current crosses zero can be estimated. Accordingto the exemplary embodiment of the present disclosure, I_(refa),I_(refb), and I_(refc), which are intermediate outputs of thefeedforward controller, correspond to the estimated values of thecurrent. The dead band compensation block 150 a and the adder 160 a canperform dead band compensation by a following equation using the output.IF I _(refn)≅0THEN V _(DUTY)+sign(I _(refn))Duty_(dead)  (21)

An IF clause in the equation (21) is not I_(refn)=0. That is, when themotor current is within a predetermined range that can be considered tobe virtually zero, then it can be regarded as “zero crossing”. In thedescription, in addition to the case where the zero crossing is actuallyperformed, the case where it can be regarded as the zero crossing isalso collectively referred to as the “zero crossing”.

In the method of the present modification, since the noise level of thesignal used for the dead band compensation is low, it is not necessaryto perform a limit cycle vibration prevention measure, and it isexpected that the dead band can be guaranteed with high accuracy.

In addition, since the timing at which the current crosses zero isestimated by using the intermediate output of the FF type controller, anestimation model and an FF type controller model can be matched witheach other. Accordingly, it is possible to reduce the order of thecontroller. Further, by independently controlling the three phases, thedead band of the drive circuit can be efficiently compensated.

When compensating for the dead band, it is not essential to provide thedisturbance observer described above. The feedforward control ispossible without providing the disturbance observer.

Next, application of the above-described embodiment and modificationwill be described.

FIG. 6 schematically illustrates a typical configuration of an EPSsystem 2000 according to an exemplary embodiment.

Vehicles such as automobiles normally have an EPS system. The EPS system2000 includes a steering system 520 and an auxiliary torque mechanism540 for generating auxiliary torque. The EPS system 2000 generates theauxiliary torque that assists steering torque of the steering systemgenerated by a driver operating a steering wheel. With the auxiliarytorque, an operation burden of the driver is reduced.

The steering system 520 includes, for example, a steering wheel 521, asteering shaft 522, universal joints 523A, 523B, a rotary shaft 524, arack and pinion mechanism 525, a rack shaft 526, right and left balljoints 552A, 552B, tie rods 527A, 527B, knuckles 528A, 528B, and rightand left steered wheels 529A, 529B.

The auxiliary torque mechanism 540 includes, for example, a steeringtorque sensor 541, an electronic control unit (ECU) for an automobile542, a motor 543, and a deceleration mechanism 544. The steering torquesensor 541 detects the steering torque in the steering system 520. TheECU 542 generates a drive signal based on a detection signal from thesteering torque sensor 541. The motor 543 generates the auxiliary torquecorresponding to the steering torque based on the drive signal. Themotor 543 transmits the generated auxiliary torque to the steeringsystem 520 via the deceleration mechanism 544.

The ECU 542 includes, for example, the controller 100, drive circuit200, and the like described above. In an automobile, an electroniccontrol system with the ECU as a core is constructed. In the EPS system2000, a motor control system is constructed by, for example, the ECU542, the motor 543, and an inverter 545. As the motor control system,the above-described motor control system 1000 can be suitably used.

The embodiment of the present disclosure is also suitably used in motorcontrol systems of an X-by-wire such as a shift-by-wire, steer-by-wire,brake-by-wire and the like, and a traction motor and the like thatrequire torque angle estimation capability. For example, the motorcontrol system according to the embodiment of the present disclosure maybe mounted on an automated vehicle that complies with a level 0 to 4(automation criterion) defined by the Japanese government and theNational Highway Traffic Safety Administration (NHTSA).

The embodiment of the present disclosure may be widely used in variousdevices including various motors such as a vacuum cleaner, a dryer, aceiling fan, a washing machine, a refrigerator, an electric powersteering device, and the like.

The invention claimed is:
 1. A controller used in a motor control systemfor driving a motor by using a drive circuit and an inverter, thecontroller comprising: a current control block, wherein feedforwardcontrol by using a current value is performed; a term of aself-inductance of the motor included in the current control block iscompensated by an inverse model; a phase characteristic of a transferfunction of the inverse model is compensated by an advance component;and a phase delay and a gain reduction of a torque output generated bythe self-inductance is compensated by correcting a gain characteristicof the transfer function of the inverse model by a function of aphysical quantity obtained based on an angular velocity of the motor. 2.The controller according to claim 1, further comprising: a disturbanceobserver for performing feedforward control by using a current value,wherein a disturbance parameter of the current control block iscompensated by adaptive control by using an output value of thedisturbance observer.
 3. The controller according to claim 1, furthercomprising: a dead band compensation block for determining a timing atwhich a dead band of the drive circuit is generated and outputting adead band compensation value; and an adder for adding a duty value atthe timing and the dead band compensation value to generate a PWMsignal.
 4. A motor control system comprising: a motor; a controlleraccording to claim 1; a drive circuit for generating a control signalfrom the PWM signal output from the controller; and an inverter forperforming a switching operation based on the control signal to allow acurrent to flow through the motor.
 5. The motor control system accordingto claim 4, wherein the motor is a motor having 10 poles and 12 slots or14 poles and 12 slots.
 6. An electric power steering system comprisingthe motor control system as claimed in claim
 4. 7. The motor controlsystem according to claim 4, further comprising a housing integrallyaccommodating the motor, the controller, the drive circuit, and theinverter.